1. Field of the Invention
The present invention is in the field of measurement of currents produced by pulses having fast rise times.
2. Description of the Related Art
Fast-pulsed power measurements and sensors to make these measurements were needed in the atomic age, where researchers needed to monitor fast high voltage signals produced by Compton diode radiation sensors located close to nuclear explosions. The Compton diode provided voltage waveforms greater than 10,000 volts which were analyzed to determine details of the nuclear chain reaction. To “freeze” the many details of the reaction requires fast accurate voltage measurements on the order of a nanosecond or less. Much high power pulse instrumentation is largely the result of the technology developed for government laboratories weapons research.
The technology developed for those measurements now finds use in the semiconductor industry. Microprocessor circuitry is incredibly complex and very sensitive to uncontrolled pulses of energy, such as static electric discharges (Electrostatic Discharge Voltages—“ESD”). Protection circuits are required on every connection which every microprocessor makes to the outside world, and these circuits must be tested. A pulse that accurately simulates ESD is applied to the protection circuit. The voltages and currents reacting with this protective circuit are monitored at pulse amplitudes up through circuit failure. The information obtained during the tests is used to assess the efficacy of each protection circuit design as well as to obtain information needed to improve future protection circuits.
High frequency RF signals or fast pulses in the time domain are usually transported on coaxial transmission lines. In most cases voltage measurement on the line (which is easily accomplished) provides a reasonably close (calculated) value for the current in the line throughout the total pulse. The accuracy of the current calculation requires the measurement of voltage values over a narrow resistance range. When using a 50-ohm coax transmission line, the greatest accuracy for the calculated current values occurs for resistance values in the immediate vicinity of 50 ohms. The testing of ESD protective circuits in semiconductors encounters resistance values that can vary between 1 and 10,000 ohms. As a consequence, obtaining accurate values for voltage and current over this entire dynamic resistance range requires that voltage values and current values both be actually measured, preferably with accuracies of approximately 1% or better.
The present environment of this current sensor relates to the generation and measurement of electrical signals that are intended to simulate electrostatic discharge (ESD) voltages across and current pulses through semiconductors. ESD pulses are regularly observed with rise times of 0.1 nanosecond or less, and having durations as long as a few hundred microseconds. Other electrical overstress threats are of a longer duration, up to 1 millisecond or more.
Additional design constraints are provided by the requirements of the expensive digitizers (similar to oscilloscopes) used to display this current and voltage data. To optimally use their capabilities requires that the sensor speed be about three times faster than the digitizer. In the ESD environment, this requires sensors capable of providing data with 20- to 30-picosecond rise times to the digitizer. Additionally, the sensors must measure pulse time durations of up to one microsecond or longer. These conflicting requirements present problems to the prior art sensors.
In a coaxial transmission line, current flows on both conductors in equal but opposite amplitudes, not just on the inner conductor as is sometimes believed. One type of current sensor, the current transformer, places the sensing element between the inner and outer conductors of the coaxial transmission line. The opposite currents flowing through the two conductors produce a magnetic field between them, which couples to a toroidal magnetic (e.g., ferrite) core placed between the two conductors. A multi-turn winding on the ferrite core provides an output voltage that is proportional to current flow in the coaxial line. However, the parasitic inductances and capacitances in the windings limit its accuracy over the time periods needed for ESD measurements.
A similar current sensor eliminates the windings and utilizes a magnetic core having a gap where the field is concentrated. A Hall-effect sensor is placed in the gap to measure the field. U.S. Pat. No. 5,583,429 to Otaka discloses such a current measuring method. Hall-effect sensors are both insensitive and slow—their fastest response time of slightly faster than one microsecond renders them unable to be used for faster pulse current measurements. Additionally, the split ring ferrite core distorts the current flowing inside the coaxial transmission line, making a meaningful measurement all the more difficult.
A still further variation is the Rogowski coil, in which the coil couples to the magnetic field without a magnetic core. The Rogowski coil is constructed as a constant impedance transmission line wound around a center conductor, which enables measurement of the fastest parts of the current. However, inherent inductive and capacitive parasitics in the windings result in an objectionable broad band resonance at the end of the pulse, which seriously limits the measurements fidelity of long pulses.
Another family of current sensors, known as “current viewing resistors,” utilizes resistors placed in the outer conductor to obtain current measurements. There are several variations. In one variation, an end-of-line sensor places a current sensing resistor in the outer conductor at the end or beginning of a transmission line where one terminal of the resistor can be grounded. The current sensing resistor is typically cylindrical and is grounded on one end. Accordingly, capacitive currents to ground on the return side, which could distort the signal propagating along the coaxial cable, are eliminated. U.S. Pat. No. 3,646,440 to Wilhelm discloses such an end-of-line current sensor resistor.
The non-constant impedance and torturous current flow path of the Wilhelm '440 sensor could not provide sub-nanosecond rise time measurement capabilities. In fact, the Wilhelm '440 description identifies the response requirements as being one microsecond. Other current viewing resistors produced by T & M Associates of Albuquerque, N. Mex., provide much faster pulse responses, including some embodiments having response capabilities of a few nanoseconds.
Very high-speed coaxial line current sensors have also utilized in-line resistors, which measure current across a short section of high impedance line inserted into the outer conductor. This current sensing method differentiates the current waveform and effectively eliminates the low frequency or slow speed time domain part of the signal. After the sensed signal is coupled to the measuring circuit, the measured signal must be integrated to recover the true current waveform at longer times. This integration essentially recovers the low frequency part of the waveform along with the high speed part; but loses the base line reference, which poses a problem when seeking accurate current measurements.
Additionally, passive high-speed integrators have limited rise time capabilities, which in turn limit the fastest rise time capability of the total measurement. The fastest passive integrators have undistorted rise time limitations of about 0.5 nanosecond. Calibration of integrators has limitations of about 5% to 10% at best, further limiting the accuracy of the integrated signal. For long time response, the time for a specific amount of droop at long time is also limited in a passive integrator by reasonable capacitor values.
U.S. Pat. No. 2,423,447 to Grimm uses a cylindrical resistor in series with the outer conductor of a coaxial line. This resistor is kept short and small (compared to the measurement wavelength) to minimize the distortion caused to the current flowing in the coaxial line. After sensing the current in the outer conductor as a voltage, the signal is then rectified to convert it to a DC signal, which achieves isolation from the current sensing resistor so that it can be taken to an external location for monitoring. A shunt capacitor shunts RF signals to ground while allowing the DC signal to pass through to the external indicating meter.
As disclosed by Grimm '447, the DC meter, which indicates RF current passing through the sensing resistor after the conversion elements change RF current to DC current, must be located close to the current source to achieve accurate measurements. If Grimm '447 had isolated the current sensing resistor with the (now) more commonly-used series isolation resistors, followed by RF blocking capacitors to the common return, the remote DC measurement could have been placed at any distance from the current sensing resistor without disturbing the signals on the coaxial transmission line.
The Grimm '447 resistor is not entirely coaxial, and thus presents some amount of impedance discontinuity with respect to the coaxial transmission line, which causes minor signal reflections at the frequencies used for this circuit. Grimm '447 also does not place a shield around the current sensing resistor and detection circuit, which will cause this circuit to be sensitive to either RF or DC noise currents, picked up on the coaxial cable that may flow through his current sensing resistor. Small amounts of DC or noise voltage present at opposite ends of a coaxial cable will cause noise currents to flow through the current sensing resistor and produce errors in the coaxial cable current measurement.
Grimm '447 converts the sensed signal into a DC signal, which allows the sensed signal to be transported a limited distance beyond the current viewing resistor without disturbing the sensing circuit or elements. To obtain accurate pulse current measurements requires that pulse current signal remain in its original condition, which maintains the original waveform parameters.
Another in-line resistor sensor is disclosed in U.S. Pat. No. 3,243,704 to Jarger et al. A cylindrical resistor in series with the outer conductor forms the current sensing element used by the RF reflectometer of Jarger et al. '704. A shielded container houses the current sensing element to protect it from noise currents that may travel on the coaxial cable shield, shunting any noise currents around the sensing resistor circuit. Like Grimm '447, Jarger et al. '704 converts the sensed signal to DC and then uses series resistor and shunt capacitor isolation to transport the DC signals out of the shielded box.
The Jarger et al. '704 reflectometer places a ferrite toroid at each side of the coaxial line where it leaves the shielded box to isolate the current sensing resistor from the ground return of the shielded container. This ferrite placement is entirely appropriate for the Jarger et al. '704 low frequency signals (approximately 30 MHz), which have an equivalent rise time of about 10 nanoseconds. The limited capacitance effects at 30 MHz resulting from the length of coaxial cables between the current sensing resistor and the ferrites placement are not likely to cause distortion of the sensed signal throughout the Jarger et al. '704 frequency range of interest. For pulse shapes having rise times that are one or more orders of magnitude faster, such capacitance effects cannot be ignored.
Fast time domain signals and extremely high RF signals present design problems not found in the low frequencies of Jarger et al. '704. For example, the Jarger et al. '704 reflectometer places a 90-degree corner in the coaxial line between the ferrites and the current sensing resistor. Additionally, the series current sensors of Grimm '447 and Jarger et al. '704 do not seek to maintain a constant impedance in the coaxial line carrying the current to be sensed, which is required to avoid generating reflections in the current sensing resistor area of the coaxial transmission line that risk changing the pulse waveform before it is measured.